Frequency responsive network



Jan. 3, 1967 w, BRAULT 3,296,463

FREQUENCY RESPONSIVE NETWORK Original Filed Oct. 21, 1963 5 Sheets-Sheet 1 ea 01/! 8 /0 I I I l W0 W0 Frequency Frequency F IN VENTOR.

James W. BI'CIUH BY 771mm 8i ATTORNEYS Jan. 3, 1967 J. w. BRAULT 3,296,463

FREQUENCY RESPONSIVE NETWORK Original Filed Oct. 21, 1963 5 Sheets-Sheet 2 INVENTOR.

- James W. BruuH ATTORNEYS United States Patent 3,296,463 FREQUENCY RESPONSIVE NETWORK James W. Branlt, Princeton, N.J., assignor to Princeton Applied Research Corporation, a corporation of New Jersey Original application Oct. 21, 1963, Ser. No. 317,718, now patent No. 3,270,213, dated Aug. 30, 1966. Divided and this application Dec. 20, 1965, Ser. No. 536,247

7 Claims. (Cl. 307-885) This application is a division of US. application Serial No. 317,718, filed Octboer 21, 1963, now US. Patent No. 3,270,213 granted August 30, 1966.

This invention relates generally to frequency responsive networks, and more particularly to tunable frequency responsive networks employing resistance-reactance circuit elements.

Two common forms of frequency selective band pass or band reject networks using only one type of reactance element, either inductive or capacitive, are known as Wein bridge or twin-T networks. To adapt frequency selective circuits of this type for tuning over a range of frequencies, it is necessary that two or more elements be varied in precise tracking relationship to maintain a desired frequency response characteristic for the network.

It is an object of the present invention to provide an improved frequency selective network, using only one type of reactance device, which is tunable over a relatively wide range of frequencies using only a single variable element, without significant variations in the frequency response characteristics of the network.

It is another object of this invention to provide an improved frequency selective network, using only a single type of reactance device, which may be tuned over a very wide range of frequencies using only two variable elements which need not be maintained in close tracking relation to maintain a desired frequency response characteristic.

A frequency selective network in accordance with the invention includes input circuit means with resistive and either capacitive or inductive circuit elements. Impedance isolation means such as a buffer amplifier is coupled between the input circuit means and a summing network including the series connection of a resistive circuit element and a reactive circuit element. Utilization circuit means is coupled between the junction of the series connected summing network elements and a point of reference potential. The isolation means causes a voltage to be applied across the summing network which is effectively equal to the difference between the resistive and reactive voltage components developed respectively across resistive and reactive elements of the input circuit.

To provide a relatively sharp ferquency response, it is desirable that the time constant of the resistive and reactive elements of the input circuit be made approximately equal to the time constant of the resistive and reactive components of the summing network. However, due to the isolation means between the summing circuit and the elements of the input circuit, it has been found that the relationship of the time constant of the elements of the input circuit means to that of the summing nework is not cirtical as in prior circuits. As a result, the network can be tuned over a wide range of frequencies by varying only one of the elements of the summing network. In tuning over a frequency range of about 4 to 1, it was found that the effective figure of merit or Q of the network changed only about 110% and accordingly that the frequency response character istic thereof was not materially altered over the entire frequency range to which the circuit was tuned.

Where the input circuit means includes a pair of time "ice constant circuits each comprising a resistive and a reactive device connected in a manner similar to that of the input circuit loops of a twin-T network, it is desirable to maintain equal time constants in the two time constant circuits to provide a relatively sharp frequency response. If a network of this type is to be used to tune a frequency range significantly greater than that mentioned above and yet maintain the desired bandpass characteristic, it is necessary that the time constants of each of the time constant circuits of the input circuit means be adjusted in fairly precise tracking relation as the time constant of the summing network is varied. On the other hand, the isolation between the summing network and the input circuit means does not require that the time constant of the summing network be precisely tracked to the time constants of either of the time constant circuits of the input circuit means.

The isolation means permits a further simplification of the frequency selective network and a relaxation of the tracking requirements of the tuning elements and at the same time permits tuning of the network over a very wide range of frequencies. A simplified circuit embodying the invention includes only a single time constant circuit including resistive and reactive circuit elements across the signal input terminals to which a signal is applied. The isolation means couples the input circuit means to the summing network as aforesaid. It was found that this circuit could be tuned over a frequency range of the order of thirty to one without materially changing the frequency response thereof by simultaneous variation of individual elements in the summing and input circuits. Since the ratio of the time constants of the summing network and the input circuit can be varied over a wide range without materially altering the frequency response characteristic, precise tracking of these elements is not necessary.

It will be understood that the frequency selective network of the invention may be used in a wide variety of applications such as, but not exclusive to amplifier circuits and oscillator circuits.

The novel features which are considered to be characteristic of this invention are set forth with particularity in the appended claim. The invention itself, however, both as to its organization and method of operation as well as to additional objects and advantages thereof will best be understood from the following specification when read in connection with the accompanying drawings in which:

FIGURE 1a is a schematic circuit diagram partially in block form of a resistance-capacitance frequency selective network embodying the invention;

FIGURES 1b and 1c are alternate resistance-capacitanc networks which can be alternatively used in the circuit of FIGURE 1a to produce respectively the frequency response characteristics shown in FIGURES 3a and 3b;

FIGURE 2 is a graph showing the relationship of the effective Q of the circuit of FIGURE 1a to the ratio of the time constant of the summing network to the time constant of the input circuit;

FIGURES 3a and 3b are graphs showing the frequency response characteristics which may be obtained with circuit of FIGURE 1a;

FIGURE 4 is a schematic circuit diagram partially in block form of a resistance-capacitance frequency selective circuit in accordance with an embodiment of the invention;

FIGURE 5 is .an equivalent circuit diagram representa tive of the circuits of FIGURES 1a and 4;

FIGURE 6 is an equivalent circuit diagram illustrating 3 a modification of the equivalent circuit diagram of FIG- URE 5;

FIGURE 7 is an equivalent circuit diagram representing a simplification of the equivalent circuit diagram of FIGURE 6;

FIGURE 8 is an equivalent circuit diagram illustrating a modification of the equivalent circuit diagram of FIG- URE 7;

FIGURE 9 is a schematic circuit diagram partly in block form representing an exemplification of the equivalent circuit diagram of FIGURE 8; and

FIGURE 10 is a detailed schematic circuit diagram of a tunable frequency selective amplifier embodying the invention.

Like reference characters will be used to identify like components in the various figures of the drawings.

Although the circuits described and shown herein comprise only resistance-capacitance frequency selective networks it will be understood that the underlying principles are also applicable to resistance-inductance networks, and the term reactive element as used in the specification and claims will be deemed to apply to either a capacitive element or to an inductive element.

The circuit of FIGURE 1a bears some resemblance to that of a twin or parallel-T network, and may be analyzed in a manner somewhat similar to that set forth by H. H. Scott, A New Type of Selective Circuit and Some Applications, Proc. I.R.E. vol. 26, pp. 226-235; February 1938, and by W. N. Tuttle, Bridged-T and Parallel-T Null Circuits for Measurements at Radio Frequencies, Proc. I.=R.E., vol. 28, pp. 23-29; January, 1940.

Basically the circuit of FIGURE It: includes an input circuit comprising a differentiator R C and an integrator R C coupled to a pair of signal input terminals 10. The time constants of the differentiator and the integrator are equal. The resultant voltages across the resistor R and capacitor C are passed respectively through impedance isolation means 12 and 14 which may be buffer amplifiers such as cathode or emitter followers having a high input impedance and a low output impedance.

The voltage output from the isolation means 12 and 14 are combined in a summing network Z and Z and an output signal may be derived between the junction of the impedance elements Z and Z and a point of reference potential shown as ground. Impedance isolation provided by the isolation means 12 and 14 provides a new degree of freedom in the design of tunable resistancereactance frequency responsive networks not found in tunable passive networks heretofore known in the art.

The open circuit transfer function for the circuit of FIGURE la is:

When it is specified that T =T =T the transfer function is:

the type shown in FIGURE 1b having corresponding terminals. In this case the impedance element Z comprises a resistor R and the impedance element 2.; comprises a capacitor C In such a case, the network of FIGURE la is an active bandpass filter whose transfer function is:

Where T ==R C In the special case where K =K =1, the frequency of resonance becomes:

The response characteristic of the network of FIGURE 1a where the impedance elements Z and Z; are resistive and capacitive respectively is shown in FIGURE 3a. From Equation 4 above it can be seen that the frequency of resonance can be varied by changing T For example, either the resistor R or the capacitor C may be varied to change the response frequency. From Equation 5 above it will be seen that a variation In T also changes the effective Q and hence the frequency bandpass response of the network.

With reference to FIGURE 2 which is a plot of network Q as a function of the ratio T /T it will be seen that T may be varied over an extended range of about siXteen-to-one with a Q variation of only :L-10%. Hence a variation of T over a range of sixteen-to-one produced only a very slight change in Q, and, therefore, only a very small change in frequency response as the network is varied over a frequency range of about four to one. It is significant that the circuit can be tuned over such a frequency range without material changes in bandpass characteristic using only a single tuning control element such as the resistor R or the capacitor C The second alternative of the circuit of FIGURE 1a I This network has a frequency null for any value of R C occur-ring when K TT In the special case when K =K =1, the notch frequency is:

And the effective figure of merit or Q of the network The response characteristic of the network as now modified is shown in FIGURE 3b, and the notch frequency can be varied by changing T From Equation 9 above it can be seen that T; can be varied over a relatively wide range without causing large changes in Q. Accordingly the network can be tuned over a relatively wide range of frequencies by adjusting only a single control element such as R; or C without causing a material change in frequency response.

In order to increase the frequency range over which the network of FIGURE 1 may be tuned while maintaining a given tolerance of Q variation, then T must be varied as well as T or T An important condition of the network is that T=T =Tg, accordingly T and T must be varied in close tracking relation or the transmission of the network at resonance deteriorates rapidly. On the other hand, due to the isolation means 12 and 14, there is no stringent requirement that T closely track T and T The actual extending tuning of the circuit of FIG- URE 1 may be effected by ganging R and R with the resistor in the summing network for unicontrol operation, if special care is taken to insure that R and R are adjusted in close tracking relation.

The frequency selective network of FIGURE 4 simplifies the extended range tuning problem by eliminating the need for two separate tuning control devices in the input circuit. This network is capable of covering a frequency range of the order of thirty to one by varying simultaneously only two resistors which need not be kept in precise tracking relation. The input circuit of the FIGURE 4 network includes a series variable resistor 20 and a shunt capacitor 22. The voltage across the resistor 20 which is the difference between the input voltage at the terminals 24, and the voltage across the capacitor 22, is applied to the unity gain ditference amplifier 26. The output voltage of the amplifier 26, which provides impedance isolation between its input and output terminals, thus corresponds to the voltage across the resistor 20.

The voltage across the capacitor 22 is applied to a unity gain buffer amplifier 28. The summing network which includes a variable resistor 30 and a capacitor 32 is connected between the difference amplifier 26 and the buffer amplifier 28. The total voltage across the summing network is a function of the difference between the voltage across the resistor 20 and the voltage across the capacitor 22 and impedance isolation is provided between the input circuit and the summing circuit by the amplifiers 26 and 28.

An output or utilization circuit is connected to a pair of output terminals 34 one of which is at ground potential, and the other of which is connected to the junction of the resistor 30 and the capacitor 32.

The frequency selective network is tuned by the simultaneous adjustment of the resistors 30 and 20 which may be ganged for unicontrol operation as indicated by the dashed line 36. Resistors 30 and 20 are adjusted to maintain the time constant of the resistor zit-capacitor 22 network approximately equal to the resistor Sit-capacitor 32 network. As noted above in connection with FIGURE 3, the ratio of the time constant of the summing network to that of the input circuit may vary over a relatively wide range without appreciably altering the Q or the frequency response characteristic of the network.

The circuit of FIGURE 4 comprises a bandpass network having a frequency response characteristic as shown in FIGURE 2a. If desired, the circuit may be altered to provide a null response characteristic as shown in FIGURE 2b by interchanging the positions of the variable resistor 30 and the capacitor 32. Alternatively a null response network may be produced by interchanging the positions of the variable resistor 20 and the capacitor 22. If desired, a bandpass characteristic may be produced by interchanging the positions of the variable resistor 30 and the capacitor 32 as well as the positions of the variable resistor 20 and the capacitor 22.

The circuits of FIGURES la and 4 may be represented by the equivalent circuit diagram of FIGURE 5. A voltage source 40 providing a voltage c is connected between a point of reference potential, shown as ground, and one end of the summing network comprising a variable resistor 42 and a capacitor 44 connected in series. A voltage source 46 providing a voltage a is connected between ground and the other end of the summing network. Utilization circuit means is represented by a load resistor 48 connected between ground and the junction of the resistor 42 with the capacitor 44.

The voltage source 40 provides a voltage e equivalent to that developed across the resistor R of FIGURE 1, or the resistor 20 of FIGURE 4. In like manner, the voltage source 46 provides a voltage e equivalent to that devel- 6 oped across the capacitor C of FIGURE 1 or the capacitor 22 of FIGURE 4.

The equivalent circuit of FIGURE 6 is a modification of that shown in FIGURE 5 wherein the voltage sources 40 and 46 are connected to ground respectively through voltage sources 50 and 52 each providing a voltage e Voltage e is the same as the voltage 2 from the source 46 except that it is shifted in phase by To balance out the effects of the additional voltage sources 50 and 52, a voltage source 54 providing a voltage +e is added to the output voltage from the summing network in an adder circuit 56. The adder circuit provides impedance isolation between the voltage source 54 and the summing network.

Since the voltage sources 46 and 52 provide equal and opposite voltages, they may be eliminated, and the terminal of the capacitor 44 to which these sources were connected may be grounded as shown in FIGURE 7. The circuit of FIGURE 7 provides an advantage over that of FIGURE 4 in that particular care must be exercised in the design of the amplifier 28 which drives the capacitor 32, so that the output resistance thereof does not adversely affect the time constant of the summing network. As shown in FIGURE 7, the capacitor 44 is grounded, and only the resistor 42 is driven from the input circuit.

The equivalent circuit diagram shown in FIGURE 7 may be redrawn by replacing the voltage sources 40 and 50 with an equivalent voltage source 58 which provides a voltage (e ,,2.e Note with respect to FIGURE 4 that the sum of the voltages across the resistor 20 (e,) and the capacitor 22 (c is equal to the signal input voltage (e Also in FIGURE 1 when T =T the sum of the voltages across the resistor R and capacitor C is equal to the signal input voltages. This relationship is expressed as:

Replacing the voltage source 40 of FIGURE 7 with its equivalent (e -e as noted in Equation 11, and combining with the voltage source 50 (e provides a resultant voltage source (e -22 as shown in FIGURE 8.

FIGURE 8 is the equivalent circuit diagram of the schematic circuit diagram, partly in block form shown in FIGURE 9. A signal input voltage e applied to a pair of input terminals 24, is developed across the series combination of a resistor 20 and capacitor 22 in the same manner as shown above in FIGURE 4. The voltage across the capacitor 22 is applied to an amplifier 60 which provides a gain of two and a phase reversal of the input signal. The output signal from the amplifier 60 (2e,,) is applied together with the input signal (e to an adder circuit 62. The output signal from the adder circuit 62 drives the summing network 42, 44 as described above.

The voltage across the capacitor 22 is also applied to the adder circuit 56 previously described in FIGURES 6-8. It should be noted that the adder circuits 56 and 62 not only serve to linearly add the signals applied thereto, but also provide impedance isolation between the input circuit and the summing circuit.

The circuit of FIGURE 9 provides a bandpass response characteristic of the type shown in FIGURE 3a. The frequency of response may be varied over an extremely wide frequency range, of the order of thirty to one, by conjointly varying the resistors 20 and 42 in approximate tracking relation. The tracking is not critical, and with relatively'wide variations between the time constants of the input and summing circuits, the Q or frequency response characteristic of the network remains substantially constant over the entire frequency range.

The frequency response characteristic of the network may be changed to provide a null by receiving the positions of the resistor and capacitor in either the input or summing circuits. If the positions of the resistor and capacitor in both the input and summing circuits are reversed, then the network again exhibits a bandpass characteristic.

At this point it should be noted with respect to the figures discussed thus far that resistance-inductance networks may also be used to provide wide range tuning. Furthermore, a resistance-capacitance network and a resistance-inductance network can be used respectively in the input and summing circuits or vice versa, to provide wide range tuning.

A practical exemplification of the block diagram circuit of FIGURE 9 is shown in the schematic circuit diagram of FIGURE 10. The circuit of FIGURE 10 comprises an amplifier which is tunable over a range of about thirty to one. The various transistor stages used in this amplifier are of conventional design, and accordingly a detailed description of all the circuit components is unnecessary.

A signal input voltage (e to be amplified is applied to a pair of input terminals 24 and is coupled through a capacitor 64 to the base electrode of a transistor 66. The transistor 66 and a transistor 68 are connected in cascode relation. The amplified input signal (e is developed in the collector circuit of the transistor 68 and applied to the base electrode of a transistor 70 which is connected as a phase splitter.

, One output signal from the phase splitter is developed across the emitter resistor of the transistor 70 and applied to a phase inverter stage including a transistor 72. The signal voltage developed at the collector electrode of the transistor 72 corresponds to, and is in phase with the applied signal voltage (e and is applied to the input circuit time constant network 74. As mentioned hereinabove with reference to FIGURES 4 and 9 the input circuit time constant network comprises a series variable resistor 20 and a shunt capacitor 22.

The voltage developed across the capacitor (e is applied to the base electrode of a transistor 76. The output signal from the transistor 76 is developed at its emitter elect-rode. The collector electrode of the transistor 76 is directly connected to the base electrode of a transistor 80. The collector electrode of the transistor 80 is connected in common to the emitter electrode of the transistor 76 to provide a feedback loop to raise the input impedance and lower the output impedance of the transistor 76. The signal at the emitter electrode of the transistor 76 is applied to the base electrode of a transistor 78, which is connected as an emitter follower.

The voltage (2 at the emitter electrode of the transistor 78 is applied to the base electrode of a transistor 82, which provides two times gain for signals applied to its base elect-rode. The signal current corresponding to (e at the collector electrode of the transistor 70 flows into the emitter electrode of the transistor 82, and a resultant output signal (s -22 is developed at the collector electrode of the transistor 82. Thus, the circuit including the transistor 82 cor-responds to the amplifier 60 and the adder circuit 62 of FIGURE 9.

The resultant voltage (e 2e at the collector electrode of the transistor 82 is applied to the summing network 84 which includes a variable resistor 42 and capacitor 44 connected in series between the collector of transistor 82 and a point of reference potential which in this case is the negative terminal of the operating potential supp y- The resultant voltage across the capacitor 44 is applied to a transistor 86 which is connected as an emitter follower. The output current from the collector electrode of transistor 86 provides base current drive for a transistor 88. The collector electrode of the transistor 88 is connected in common to the emitter electrode of the transistor 86 to provide a feedback loop to raise the input impedance and lower the output impedance of the transistor 86.

The collector electrode of the transistor 90 and the collector electrode of the transistor 78 are connected in common to provide an adder circuit corresponding to the 8 adder circuit 56 of FIGURE 9. In order to maintain the circuit in balance, the gain from the base electrode of the transistor 86 to the collector electrode of the tran sistor 92 matches the gain from the base electrode of the transistor 76 to the collector electrode of the transistor 92.

The signal output voltage from the adder circuit is taken from the collector electrode of the transistor 92 and is applied to a phase splitter including a transistor 94. The output terminals from the network comprise either of the terminals 96 or 98 and ground.

A regenerative feedback network is provided from the emitter electrodes of the transistors and 88 to the collector electrode of the transistor 66 to enhance the overall frequency response of the network. In this respect it will be noted that the transistors 80 and 88 are connected to add the currents through the emitter resistors of the transistors 76 and 86 so as to produce a resultant feedback signal corresponding to the output signal from the network, but isolated therefrom.

The amplifier of FIGURE 10 is tuned by varying the resistors 2t} and 42 in rough tracking relation. If desired, the resistors may be ganged for unicontrol operation as indicated by the dashed line 10! The particular band of frequencies over which the amplifier is tunable is a function of the time constants of the input and summing circuits. A practical example of an amplifier provides a tuning range of 1.5 c.p.s. to k-c. in five bands by simultaneously switching dilferent capacitors in place of the capacitors 22 and 44 when going from one band to another. The time constants of the input and summing networks are kept equal, but as described above considerable latitude in this relationship can be tolerated without affecting the Q of the network. As a result, the troublesome and expensive requirement of maintaining close tracking relation between the resistors 20 and 42 is eliminated.

Of considerable importance is the fact that circuits embodying the invention maintain the transmission characteristic of the network at resonance constant as the circuit is tuned over its wide frequency range. The reason that this is important in the circuit of FIGURE 10 is that the feedback loop magnifies any changes in the transmission characteristic of the network and thereby seriously changes the gain and selectivity of the overall circuit.

What is claimed is:

1. A tunable frequency selective network comprising in combination,

input circuit means including resistive and reactive circuit elements connected in series between a signal input terminal and a point of reference potential for said network,

summing circuit means including resistive and reactive circuit elements,

the time constant of said summing circuit means being substantially equal to the time constant of said input circuit means,

isolation circuit means coupled between said input circuit means and said summing circuit means to apply to said summing circuit means a voltage which is effectively equal to the difference in voltage appearing across resistive and reactive circuit elements of said input circuit means,

output circuit means coupled between said summing circuit means and said point of reference potential, and

means for adjusting the value of one of said circuit elements of said summing circuit means to tune said frequency selective network.

2. A tunable frequency selective network comprising in combination,

input circuit means including resistive and reactive circuit elements connected in series between a signal input terminal and a point of reference potential for said network,

9 summing circuit means including resistive and reactive circuit elements connected in series, the time constant of said summing circuit means being substantially equal to the time constant of said input circuit means, isolation circuit means coupled between said input circuit means and said summing circuit means to apply across said summing circuit means a voltage which is a function of the difference in voltage appearing across resistive and reactive circuit elements of said input circuit means,

output circuit means coupled between a point of reference potential for said network and the junction of the resistive and reactive circuit elements of said summing circuit, and

means for adjusting the value of one of said circuit elements of said summing circuit means to tune said frequency selective network.

3. A tunable frequency selective network comprising in combination,

means providing a pair of signal input terminals,

input circuit means including resistive and reactive circuit elements connected in series across said input terminals,

summing circuit means including resistive and reactive circuit means connected in series, the time constant of said summing circuit means being substantially equal to the time constant of said input circuit means,

an isolation amplifier having an input circuit coupled across one of said resistive and reactive circuit elements and an output terminal,

a differencing amplifier having an input circuit coupled to receive the input signal voltage from said input terminals and the voltage applied to the input circuit of said isolation amplifier and an output terminal,

means coupling said summing network between the output terminals of said isolation amplifier and said differencing amplifier,

output circuit means coupled to the junction of the resistive and reactive circuit eelments of said summing circuit means, and

means for adjusting the value of one of the circuit elements of said summing circuit means to tune said frequency selective network. 4. A tunable frequency selective network as defined in claim 3 inluding means for adjusting the value of one of the circuit elements of said input circuit means to maintain the time constant of said input circuit means substantially equal to the time constant of said summing circuit means.

5. A tunable frequency selective network as defined in claim 4 wherein the resistive circuit elements of said input circuit means and said summing circuit means are adjustable, and ganged for unicontrol operation to tune said frequency selective network.

6. A frequency selective network comprising in combination,

input circuit means including resistive and reactive circuit elements connected in series between a signal input terminal and a point of reference potential for said network,

summing circuit means including resistive and reactive circuit elements, the time constant of said summing circuit means being substantially equal to the time constant of said input circuit means,

isolation circuit means coupled between said input circuit means and said summing circuit means to apply to said summing circuit means a voltage which is effectively equal to the difference in voltage appearing across resistive and reactive circuit elements of said input circuit means, and

output circuit means coupled to said summing circuit means.

7. A frequency selective network as defined in claim 6 wherein said reactive circuit elements comprise capacitors and said input circuit means comprises a single time constant network.

No references cited.

ARTHUR GAUSS, Primary Examiner.

D. FORRER, Assistant Examiner.

UNITED STATES PATENT OFFICE CERTIFICATE OF CORRECTION Patent No. 3,296,463 January 3,1967

James W. Brault It is hereby certified that error appears in the above numbered patent requiring correction and that the said Letters Patent should read as corrected below.

Column 1, line 62, for "cirtical" read critical column 4, line 16, for "response" read resonance column 4, lines 57 to 59, equation (9) should appear as shown below instead of as in the patent:

column 6 line 29 for "(e 2e read (e -2e line 36, for "e read ecolumn 9, line 39, for "eelements" read elements column 10, line 4, for "inluding" read including Signed and sealed this 18th day of June 1968.

(SEAL) Attest:

EDWARD M.FLETCHER,JR. EDWARD J. BRENNER Attesting Officer Commissioner 0F Patenfc 

3. A TUNABLE FREQUENCY SELECTIVE NETWORK COMPRISING IN COMBINATION, MEANS PROVIDING A PAIR OF SIGNAL INPUT TERMINALS, INPUT CIRCUIT MEANS INCLUDING RESISTIVE AND REACTIVE CIRCUIT ELEMENTS CONNECTED IN SERIES ACROSS SAID INPUT TERMINALS, SUMMING CIRCUIT MEANS INCLUDING RESISTIVE AND REACTIVE CIRCUIT MEANS CONNECTED IN SERIES, THE TIME CONSTANT OF SAID SUMMING CIRCUIT MEANS BEING SUBSTANTIALLY EQUAL TO THE TIME CONSTANT OF SAID INPUT CIRCUIT MEANS, AN ISOLATION AMPLIFIER HAVING AN INPUT CIRCUIT COUPLED ACROSS ONE OF SAID RESISTIVE AND REACTIVE CIRCUIT ELEMENTS AND AN OUTPUT TERMINAL, A DIFFERENCING AMPLIFIER HAVING AN INPUT CIRCUIT COUPLED TO RECEIVE THE INPUT SIGNAL VOLTAGE FROM SAID INPUT TERMINALS AND THE VOLTAGE APPLIED TO THE INPUT CIRCUIT OF SAID ISOLATION AMPLIFIER AND AN OUTPUT TERMINAL, 